Recently, surface light sources are being more widely, not only for advertisement and personal computer displays, but also for liquid crystal display television sets and the like.
There is a demand of decreasing the size of an inverter circuit for driving these surface light sources and increasing conversion efficiency. Hereinafter, a description is given for the relation between the recent developments of inverter circuits for cold cathode fluorescent lamps and the invention disclosed in Japanese Patent No. 2733817 (U.S. Pat. No. 5,495,405).
Conventionally, the collector resonant circuit shown in FIG. 19 has been widely used as a classical type of inverter circuit for a cold cathode fluorescent lamp. This is sometimes referred to as a “Royer circuit”. However, the Royer circuit is officially defined as a circuit which reverses a switching operation by saturating the transformer. A circuit which performs the reverse operation by using the resonance on the collector side is referred to as “collector resonant circuit” or “collector resonant Royer circuit” as distinguished from the Royer circuit.
A type of inverter circuit for a cold cathode fluorescent lamp, which never uses the resonating method on the secondary side of the circuit, uses the so-called closed magnetic circuit type transformer, which has a small leakage inductance, for the step-up transformer. Under these circumstances, those skilled in the art understand that the so-called closed magnetic circuit type transformer is a transformer that has small leakage inductance. It is advantageous to make the leakage inductance of the step-up transformer in the inverter circuit as small as possible because the leakage inductance causes the output voltage on the secondary side of the transformer to drop.
As a result, under these circumstances, the resonance frequency of the circuit on the secondary side of the transformer is set at a frequency much higher than the operational frequency of the inverter circuit in order not to influence the operational frequency of the inverter circuit. Furthermore, the ballast capacitor Cb is essential to stabilize the lamp current.
Next, FIG. 20 illustrates a type of inverter circuit for a cold cathode fluorescent lamp, which is disclosed in Japanese Laid-Open Patent Publication (Kokai) No. Hei 07-211472. In this type of configuration, the resonance frequency of the circuit on the secondary side is three times as high as the oscillation frequency of the primary side circuit, as shown in FIG. 21. Accordingly, this widely used type of inverter circuit is referred to as the triple resonant circuit. In this case, the leakage inductance of the step-up transformer is favorably made larger to some extent.
As shown in the explanatory diagram as FIG. 22, the oscillation frequency and third-order harmonic of the inverter circuit in FIGS. 20 and 21 are combined to produce a trapezoidal waveform.
The actual current which flows through the cold cathode fluorescent lamp of the triple resonant circuit is represented by the waveform shown in FIG. 23.
The name of the step-up transformer illustrated in the circuit of FIGS. 20 and 21 this case has not been fixed yet. There has been debate about whether or not it should be referred to as a “closed magnetic transformer”, which is term used among those skilled in the art. Thus, the name remains uncertain. The problem of how to describe the state in which a larger amount of flux leaks, even though the magnetic circuit structure is closed, has been discussed.
The shape of the transformer used in the so-called triple resonant circuit is flat as shown in FIG. 24. In this transformer, the flux leakage is considerably larger than in a conventional one, although the magnetic circuit structure is closed. Specifically, the transformer has a large leakage inductance.
In any case, the technical idea behind the circuit of FIG. 20 increases the leakage inductance of the step-up transformer to some extent so as to form a resonant circuit between the leakage inductance and the capacitive component Cs formed on the secondary side of the step-up transformer. Also, the resonance frequency is set at a frequency three times as high as the operational frequency of the inverter circuit so as to produce the third-order harmonic in the secondary side circuit (see FIG. 21), thereby making the lamp current waveform trapezoidal (see FIG. 22). In this case, a ballast capacitor C2, which is the ballast capacitor, operates as a part of the resonance capacitor.
As disclosed in Japanese Laid-Open Patent Publication (Kokai) No. Hei 07-211472, this technical idea considerably improves the conversion efficiency of the inverter circuit and furthermore makes the inverter circuit smaller than the step-up transformer. Also, recent and current implementations of a collector resonant inverter circuit for a cold cathode fluorescent lamp is based on the technical idea of the triple resonance, and it would not be an exaggeration to say that the technique is employed in most of the collector resonant inverter circuits which are currently used.
Next, the invention disclosed in Japanese Patent No. 2733817 (U.S. Pat. No. 5,495,405), on which the present invention is based, makes the step-up transformer further smaller and to improve conversion efficiency drastically. The invention of Japanese Patent No. 2733817 helps decrease the size of the inverter circuit in a laptop personal computer and improves conversion efficiency. Particularly, it increases both the step-up transformer leakage inductance further and the capacitive component in the secondary side circuit larger at the same time in the triple resonant circuit. This results in the operational frequency of the inverter circuit to almost coincide with the resonance frequency in the secondary side circuit.
The technique utilizes an effect in which the exciting current flowing through the primary winding of the step-up transformer decreases when the inverter circuit operates at a frequency close to the resonance frequency in the secondary side circuit, thereby improving the power factor as seen from the primary winding side of the transformer and reducing the copper loss of the step-up transformer.
At the time the invention of Japanese Patent No. 2733817 was disclosed, many kinds of driving methods for the primary circuits have been used in connection with the conventional collector resonant circuit. One of these driving methods include a fixed frequency and zero current switching type driving method for performing switching by detecting the zero current through the primary side windings. Each of these series of peripheral techniques is related to the invention, and the usage of the resonance technique of the secondary side circuit in the invention.
Considering the history of changes in inverter circuits for a cold cathode fluorescent lamp, from the viewpoint of the leakage inductance of the step-up transformer, it can be seen that the step-up transformer leakage inductance increases, while at the same time the secondary side circuit resonance frequency decreases, as each new generation of the inverter circuit comes to the forefront, as shown in FIG. 25.
It should be noted that FIG. 25 is an explanatory diagram illustrating the changing relationship between the drive frequency f0 of the inverter circuit and the resonance frequency fr in the secondary side circuit with each new generation.
Improving the step-up transformer and appropriately selecting the drive frequency thereof achieve the objectives of miniaturizing the inverter circuit and improving the conversion efficiency of the inverter circuit. Regarding this matter, FIG. 26 is an explanatory diagram illustrating the improved conversion frequency obtained by the invention disclosed in Japanese Laid-Open Patent Publication No. 2003-168585 by the inventor of the present invention (U.S. Pat. No. 6,774,580-B2). Particularly, the diagram of FIG. 26 illustrates a scheme for improving the power factor as seen from the driving side, in which the horizontal axis indicates frequency, and θ indicates the phase difference between the voltage and current of the primary winding of the step-up transformer, showing that power factor is improved as θ becomes closer to zero. This scheme for promoting such conversion efficiency as seen from the driving side is disclosed in more detail below.
On the contrary, as shown in U.S. Pat. No. 6,114,814-B1 and Japanese Laid-Open Patent Publication No. Sho 59-032370, those skilled in the art consistently have advocated the technical idea that a high conversion efficiency inverter circuit is achieved by the zero current switching method.
This technical idea, however, does not consider the improvement power factor effect of the step-up transformer. Therefore, this technical idea is incorrect because the high efficiency is actually due to the reduction of heat generated in the switching transistor.
The reason will be described in detail below.
The zero current switching method is one power control method of the inverter circuit. A typical example thereof is a zero current switching type circuit as shown in FIG. 27, which is disclosed in U.S. Pat. No. 6,114,814-B1 and Japanese Laid-Open Patent Publication No. Sho 59-032370. The inventor of the present invention also discloses a similar technique in Japanese Laid-Open Patent Publication No. Hei 08-288080. The technique is described based on the U.S. Pat. No. 6,114,814-B1 as follows.
U.S. Pat. No. 6,114,814-B1 shows explanatory diagrams illustrating the operation of the conventional zero current switching type circuit shown in FIG. 11 of that patent (which is shown as FIG. 28 in the present specification), wherein A, B show a case in which no power control is performed; C, D a case in which power control is performed; E, F a case in which zero current switching operation is tried in a state that a voltage effective value advances in phase with respect to a current effective value. Also, in FIG. 12 of the abovementioned U.S. Pat. No. 6,114,814-B1 is shown as FIG. 29 of the present specification, wherein G, H show one exemplary control which is not zero current switching operation.
In FIG. 28, A shows the voltage of the primary winding of the transformer when drive power is at maximum, and B shows the current flowing through the transformer primary winding in this case. When the zero current switching method is used, the timing at which the current becomes zero is detected so as to switch on the driving means. When power is at maximum, specifically when no power control is performed, the duty ratio is at 100% and there is no phase difference between the effective value of the voltage phase and effective value of the current phase supplied to the transformer primary winding. In this condition the power factor is favorable.
Next, C of FIG. 28 shows the voltage across the transformer primary winding when the duty ratio (circulation angle) is decreased so as to control drive power, and D shows the current flowing through the transformer primary winding in this case. According to C and D of FIG. 28, the switching transistor of the driving method is turned on at the timing at which the current becomes zero. On the contrary, it is not at zero current timing when the switching transistor is turned off. In this case, there is a phase difference between the effective value phase of the voltage applied to the transformer primary winding and the phase of the current flowing through the transformer primary winding. As a result, the power factor is unfavorable in this case.
In FIG. 29, G shows a case in which power is controlled at a similarly limited duty ratio so that the effective value phase of the voltage across the transformer primary winding is in phase with the phase of the current flowing through the transformer primary winding, without employing the zero current switching method. In this case, the power factor is actually favorable as seen from the transformer primary winding side and the heat generated in the step-up transformer is small. However, this is not the result of the use of the zero current switching method.
Here, the technical idea that the zero current switching method makes higher conversion efficiency of the inverter circuit is contradicted by a discovery of the inventor of the present invention. In the technical idea of the invention disclosed in U.S. Pat. No. 6,114,814-B1, zero current switching method is eliminated in the state shown in G, H of FIG. 29, resulting in an increase in the conversion efficiency of the inverter circuit.
It should be noted that in E, F of FIG. 29 are explanatory diagrams illustrating a case in which zero current switching operation is tried in a state that a voltage effective value advances in phase with respect to a current effective value, and G, H of FIG. 29 are explanatory diagrams showing one exemplary type of control which is not zero current switching operation.
According to the comparative experiments conducted by the inventor of the present invention, however, the inverter circuits obtain clearly higher conversion efficiency by the control method of G, H of FIG. 29 than by the control method of C, D of FIG. 28.
Consequently, the theory that the zero current switching method makes the inverter circuit achieve a higher conversion efficiency is wrong.
The background against which such a misunderstanding has occurred is as follows.
Using the zero current switching method only when no power control is performed, there is necessarily no phase difference between the voltage phase and current phase of the primary winding of the step-up transformer. Therefore, the power factor of the step-up transformer is improved; the current flowing through the transformer primary winding decreases; and the current flowing through the switching transistor also decreases to a minimum. As a result, the heat generated in the step-up transformer primary winding and the heat generated in the switching transistor decrease, thereby improving the conversion efficiency of the inverter circuit. This is taken, by mistake, to mean that the zero current switching method brings high efficiency.
In the state shown as FIGS. 11A and 11B in U.S. Pat. No. 6,114,814-B1, during which no power control is performed, the operational state thereof is equivalent to the standard current-mode resonant operational state. Specifically, it is not the zero current switching method, but rather the conventional current resonant type circuit that brings the inverter circuit high efficiency.
A current-mode resonant inverter circuit is known for lighting a hot cathode fluorescent lamp. For example, the circuit shown in FIG. 30 is generally used. In such a current-mode resonant circuit, no dimmer means are provided in its basic circuit structure. Thus, when the light output is controlled in the current-mode resonant circuit, a DC-DC converter circuit is provided at a preceding stage.
FIG. 31 is an exemplary dimmer circuit of an inverter circuit for a cold cathode fluorescent lamp which combines a conventional current-mode resonant circuit, a DC-DC converter circuit at a preceding stage thereof, and the leakage flux transformer invented by the same inventor of the present invention (hereafter “the present inventor”). In this example, the DC-DC converter circuit comprises a transistor Qs, an inductance Lc, a diode Ds, and capacitor Cv.
A scheme of improving the current-mode resonant circuit itself for light control has also been proposed. FIG. 32 shows the dimmer circuit previously disclosed by the present inventor in Japanese Laid-Open Patent Publication No. Hei 08-288080, in which, in a prescribed period of time after timer circuits 10, 11 detect zero current, a frequency control circuit 12 turns off switching elements 2, 3. The timer circuits 10, 11, which are RS flip-flops, are set at zero current and reset after a prescribed period of time. In this scheme, light is controlled by the method in which, after the switching means is turned on by detecting zero current, the switching means is turned off.
A similar scheme is disclosed in FIG. 9 in U.S. Pat. No. 6,114,814-B1. That is the circuit diagram shown in FIG. 33, in which an RS flip-flop 172 is set at zero current and reset after a prescribed period of time. Both in U.S. Pat. No. 6,114,814-B1 and in Japanese Laid-Open Patent Publication No. Hei 08-288080, zero current is detected so as to turn on the switching means and to set the RS flip-flop at the same time, followed by resetting after a prescribed period of time so as to turn off the switching means. Both provide a dimmer function to the switching means in the current-mode resonant circuit, characterized in that the current delays in phase with respect to the voltage effective value when controlling light. They are completely the same technical ideas and their achievement methods are almost the same.
The present inventor himself has confirmed that, if light is controlled based on the invention disclosed in Japanese Laid-Open Patent Publication No. Hei 08-288080, when a cold cathode fluorescent lamp or hot cathode fluorescent lamp is controlled so as to be considerably dim, a larger current flows through the transistor of the switching means thereby generating heat.
In either case, since high efficiency in the inverter circuit is clearly due to the current-mode resonant type, the present inventor has disclosed the current-mode resonant inverter circuit for a discharge lamp as FIG. 34 in Japanese Laid-Open Patent Publication No. 2004-318059 (invented by the present inventor).
The major conventional current-mode resonant circuit is the half-bridge type, and at the same time, for current detecting means, a current transformer is provided immediately after the half-bridge output so as to detect current. This is known as a lighting device for a hot cathode fluorescent lamp. FIG. 35 shows one example of the inverter circuit applied in order to light a cold cathode fluorescent lamp.
FIG. 36 shows the voltage applied to the primary winding of the step-up transformer when driving a step-up circuit for a cold cathode fluorescent lamp in a conventional current-mode resonant circuit. FIG. 36 is an explanatory diagram illustrating the state of the voltage and current of the step-up transformer primary winding when driving the step-up transformer by the conventional current-mode resonant circuit. The voltage at the half-bridge output stage is applied to the step-up transformer primary winding without change. This voltage is set to VT1. In this case, VT1 creates a rectangular waveform. The current flowing through the step-up transformer primary winding is set to IT1. Switching transistors Q1, Q2 are turned on/off depending on the phase of IT1.
Next, to provide power control function for the current-mode resonant circuit current, a circuit called “zero current switching circuit” is available. However, when power is controlled by the zero current switching method disclosed in Japanese Laid-Open Patent Publication No. Sho 59-032370 so as to control a cold cathode fluorescent lamp, the power factor is not very favorable. Furthermore, since the half-bridge configuration cannot respond to low supply voltage, it is difficult to take full advantage of the power factor improvement effect disclosed in Japanese Patent No. 2733817 (U.S. Pat. No. 5,495,405).
Power factor becomes worse when power is controlled by the zero current switching circuit for the following reason.
In the conventional zero current switching circuit shown in FIG. 33, the relation between the voltage and current given to the primary winding of step-up transformer is exemplarily shown in FIG. 37. The current on the primary winding side does not create such an exemplary sine wave in practice. The voltage waveform rises by detecting the zero point of the current. The ON timing of the switching means is at zero current, but the OFF timing thereof is not at zero current.
The voltage waveform converted into the effective value is shown with a broken line. As can be seen from FIG. 37, the current delays in phase with respect to the voltage effective value. This means that the power factor is poor. With the zero current switching circuit, idle current (reactive current) increases when power is controlled, thereby increasing copper loss in the step-up transformer primary winding, so that the conversion efficiency of the inverter circuit becomes worse.
Next, a description is given for the reason why the power factor decreases using the zero current switching method with reference to monographs. When using the zero current switching method, power factor is poor particularly at a narrower duty ratio as shown in FIG. 38. This is because the current is considerably delayed in phase with respect to the voltage.
A description is given in further detail as follows.
FIG. 39 shows the relation between delay angle and circulation angle (duty ratio), as to how considerably the current waveform delays in phase with respect to the voltage effective value waveform, which is a simple inverse proportional relation.
FIG. 39 calculates how the voltage effective value phase and the current phase change along with a change in duty ratio. It is shown, for example, when the duty ratio is 25%, the delay angle of the current with respect to the voltage is 67.5 deg. From FIG. 39, the phase delay of the current with respect to the voltage when the duty ratio (duty ratio) is set at 25% can be obtained as about 67.5 deg.
As shown in FIG. 40, in the zero current switching circuit, the intersection of the frequency corresponding to the delay angle and the phase characteristic becomes the operational frequency of the inverter circuit. In the zero current switching circuit, therefore, the operational frequency deviation is unavoidable when power is controlled.
Next, consideration is given for power factor in FIG. 41 and FIG. 42.
In FIG. 41, if the load current converted on the primary side is set to a, the exciting current is represented by tan θ, and the current through the transformer primary winding is represented by 1/cos θ (reciprocal of power factor).
FIG. 42 is an explanatory diagram showing the relation among the load current converted on the transformer primary side, the exciting current, and the current through the transformer primary winding for considering power factor. FIG. 42 illustrates that a large delay angle allows a larger exciting current thereby increasing idle current.
In FIG. 42, the combined current ratio represents 1/cos θ (reciprocal of power factor). Taking the current delay in phase with respect to the voltage effective value as a current delay angle θ, the figure shows its relation with 1/cos θ (reciprocal of power factor). How much larger the current flowing through the transformer primary winding is than the load current is considered in FIG. 42 as follows. If the current delays by 67.5 deg. in phase with respect to the voltage effective value, 2.61-times larger current flows through the transformer primary winding than in a case in which there is no delay. Consequently, the power factor becomes extremely worse, and more heat is generated in the transformer primary winding due to increase in copper loss. Also, for the same reason, more heat is generated in the transistor of the switching means.
Specifically, when power is controlled using the zero current switching method, if using the duty ratio control method disclosed in each of U.S. Pat. No. 6,114,814-B1, Japanese Laid-Open Patent Publication No. Hei 08-288080 and Japanese Laid-Open Patent Publication No. Sho 59-032370 for power control, the following conclusion is obtained from a viewpoint of improving power factor.
In a state that the duty ratio is large, specifically, in a state that the current slightly delays in phase with respect to the voltage effective value, the conversion efficiency of the inverter circuit is favorable. However, when the duty ratio is small, there are long current delays in phase and consequently, the power factor becomes worse, and a larger current flowing through the transformer primary winding makes the inverter circuit conversion efficiency worse. Particularly, as the duty ratio becomes smaller thereby delaying the current in phase closer to 90 deg., idle current increases rapidly thereby making the efficiency worse significantly.
Specifically, in such a state, when the zero current switching method is applied to a laptop personal computer, if an AC adapter is used, the supply voltage becomes highest. Under these conditions, when power is restricted so as to make a liquid crystal display panel darker or the like, the current delays longest in phase. In this case, significant heat is generated in the inverter circuit in practice.
Furthermore, there is also a problem that the operational frequency deviation of the inverter circuit is unavoidable when current is controlled by the zero current switching method.
What is clear is that it is not always necessary to implement power control according to the technical idea of zero current switching in order to achieve a high-efficiency inverter circuit. On the contrary, the idea is damaging. In order to configure an inverter circuit with good conversion efficiency, the above technical idea has to be eliminated and a method which achieves the best power factor in the step-up transformer primary winding has to be applied.
As driving means for carrying out the technical subject matter described in Japanese Patent No. 2733817 (U.S. Pat. No. 5,495,405), separately excited driving means are often employed with the fixed frequency oscillation circuit composed of a capacitance C and a resistor R as an oscillation circuit. In this case, however, there are sometimes fluctuations in parasitic capacitances caused by assembly methods for mass production, thereby deviating the secondary side circuit resonance frequency. Alternatively, there are sometimes fluctuations in component values thereby deviating the drive frequency of the drive circuit on the primary side. In such situations, constant driving at the optimum resonance frequency at which the power factor is improved is difficult.
If the resonance frequency of the secondary side circuit is shifted away from the drive frequency of the primary side circuit, the efficiency of the inverter circuit becomes extremely worse. Therefore, when using fixed-frequency separately-excited driving means, the Q value of the resonant circuit of the secondary side circuit is lowered so as to obtain broad resonance characteristics thereby responding to frequency deviation. For such a reason, it is difficult to raise the Q value of the secondary side resonant circuit in the fixed-frequency separately-excited driving means.
When trying to drive the secondary side resonant circuit with a low Q value by a conventional current-mode resonant circuit, continuous oscillation becomes difficult. Therefore, consideration has to be given so as not to make the Q value too low when driving by the current-mode resonant type.
However, in a step-up transformer for a general cold cathode fluorescent lamp, the Q value of the secondary side resonant circuit is never set to high. Specifically, it is because the technical idea of setting the Q value to high is not known among those skilled in the art at the time of filing for the application of the present invention.
Consequently, in order to respond to a commercial step-up transformer for fixed-frequency drive, the value of the coupling capacitor Cc on the primary side is decreased so as to resonate with the leakage inductance of the step-up transformer on the primary winding side, thereby making the coupling capacitor Cc involved in the resonance to ensure continuous oscillation with stability. However, the measures involve problems that heat is generated easily in the step-up transformer.